Active array antennas have individual power amplifiers on each antenna branch or bundled antenna branches (sub-arrays). Although individually linearized towards its own transmitted signal, coupling between individual antenna elements in the array may cause backwards intermodulation. That is, an adjacent antenna element in the array may couple power back to the input of its neighbor antenna element amplifier, causing intermodulation and giving rise to unwanted spurious emissions. These unwanted spurious emissions can be much higher than the linearized amplifier output emissions would have been themselves. The reason for this is that the leakage signal power to its neighbor antenna element, although lower in level relative to the intentionally transmitted signal (usually 10-20 dB lower), does not undergo linearization in the neighbor signal path, but will rather form intermodulation by itself in the amplifier. Thus, there is a need to prevent intermodulation from occurring in active antenna arrays having individual power amplifiers on each antenna branch or bundled antenna branch.
Existing solutions for counteracting this unwanted backwards intermodulation behavior have certain deficiencies. These solutions usually show low improvements, and can be complicated to implement. For example, one proposed solution involves cross coupling of non-linear parts of the individual signal paths to form a compensation tree. This approach becomes problematic, however, because the number of antenna branches in an active antenna array can be quite large. Thus, the complexity increases very rapidly as more coupled antenna branches are added, and one wishes to counteract higher orders of intermodulation (i.e., not only the basic 3rd order intermodulation, but also the 5th, 7th and maybe also the 9th order intermodulation). In addition, there might also be a need to counteract memory effects in the amplifiers.
FIG. 1 illustrates an example of a passive antenna array system 100. Passive antenna array system 100 includes baseband processing unit 110, power amplifier 120, power combiner/divider and phase shifter 130, and one or more antennas 140. In passive antenna array system 100, the baseband signals from baseband processing unit 110 are boosted by power amplifier 120, which is connected to the antennas 140 by long feedback cables 150. The use of long feedback cables 150 may result in cable losses, potentially leading to decreased performance and increased energy consumption. Furthermore, installation of passive antenna systems may be more complex, and may require more equipment space.
FIG. 2 illustrates an active-array-antenna system 200. Active-array-antenna system (AAS) 200 may include radio frequency (RF) components, such as power amplifiers and transceivers integrated with an array of antenna elements. For example, AAS 200 may include baseband processing unit 210, radio transceiver array 220, and antennas 230. Baseband processing unit 210 may perform the processing functions of the AAS. Radio transceiver array 220 may include any suitable number of transceivers. Transceivers of radio transceiver array 220 may contain transmit chains and receive chains. Transmit chains may contain typical components such as filters, mixers, power amplifiers (PAs), and any other suitable components. Receive chains may contain typical components such as filtering, low noise amplifiers (LNAs), and any other suitable components. In some cases, the number of transmitters may not be equal to the number of receivers. AAS 200 may include any suitable number of antenna elements 230 in any suitable arrangement. For example, there exist a number of potential physical arrangements, which may include (but are not limited to) uniform linear, matrix and circular. Typically, cross polarized arrangements are deployed with an antenna element for each polarization. AAS 200 offers several benefits compared to deployments having passive antennas connected to transceivers through long feeder cables, such as passive antenna array 100 illustrated in FIG. 1. For example, by using active antenna array 200, cable losses may be reduced, leading to improved performance and reduced energy consumption. As another example, the installation may be simplified, and the required equipment space may be reduced.
AAS 200 may have numerous applications. As one example, AAS 200 may be able to perform one or more of cell specific beamforming, user specific beamforming, vertical sectorization, massive multiple input multiple output (MIMO), and elevation beamforming. AAS 200 may also be an enabler for further-advanced antenna concepts, such as deploying large numbers of MIMO antenna elements at an eNodeB. These techniques, however, will be useful in practice only if proper specification of relevant RF and electro-magnetic compatibility (EMC) requirements are in place. For these reasons, 3GPP started a study item, and subsequently a work item, to define these requirements as well as the corresponding test methodology. In addition, currently, 3GPP is studying full-dimensional MIMO (FD-MIMO), and the feasibility to increase the number of transmit antennas to 16/32/64 for various purposes.
In general, for active antenna systems such as AAS 200 described above, the power amplifier needs to be operated in the non-linear region to achieve good efficiency. FIG. 3 shows a typical AM/AM curve for a power amplifier. Graph 300 charts a normalized input magnitude before power amplification on the x-axis and a normalized output magnitude after power amplification on the y-axis. From graph 300, it can be seen that the input/output curve is highly non-linear. When the power amplifier operates in the non-linear region, some of the signals are leaked to the other frequency bands (i.e., adjacent carrier bandwidths).
FIG. 4 shows spectral regrowth due to power amplifier non-linearity. Graph 400 includes power spectral density plots for non-linear power amplification 405 and ideal power amplification 410. It can be seen from FIG. 4 that the power spectral density plot is distorted, and there is a leakage of the desired signal to the adjacent channel bandwidths.
Adjacent channel leakage ratio (ALCR) is used as a metric to measure the leakage due to non-linear power amplification. In FIG. 4, the ACLR with ideal power amplification 410 is around −100 dBc, while with realistic power amplification (with non-linearity 405) the ACLR is around −38 dBc. One method to compensate for the non-linearity of the power amplifier is to distort the input signal to the power amplifier such that the output signal from the power amplifier is transformed to be close to what it would have been if the power amplifier would have been linear. This is how digital pre-distortion techniques (DPD) operate.
FIG. 5 shows spectral regrowth with DPD. Graph 500 illustrates the power spectral density plots for non-linear power amplification 505, ideal power amplification 510, and power amplification with DPD 515. It can be seen from FIG. 5 that the spectral regrowth is reduced when DPD techniques are applied. ACLR in this case is around −100 dBc.
In active antenna systems, in addition to the non-linear power amplifier, the signals from the adjacent antenna elements will be leaked in the backward direction. These leaked signals pass through the non-linear power amplifier without passing through the DPD techniques, and corrupt the desired signal. This is referred to as mutual coupling or crosstalk between antenna elements. The impact due to cross talk is severe when the distance between the antenna elements is very small. The normal distance between array elements in an array is in the order of 0.5λ, which usually means a relatively strong coupling.
FIG. 6 shows spectral regrowth due to mutual coupling in addition to the non-linear power amplification per each antenna element. More particularly, graph 600 illustrates power spectral density plots for non-linear power amplification without DPD 605, ideal power amplification 610, non-linear power amplification with DPD 615, power amplification and mutual coupling with DPD 620, and power amplification and mutual coupling without DPD 625. In FIG. 6, the value of mutual coupling was set to −18 dB between the antenna elements. It can be observed that when antenna coupling is present between antenna elements, DPD techniques fail, as the ACLR is increased to −38.2 dBc. This implies that it does not meet the requirements as set by 3GPP.
To mitigate the impact of mutual coupling, the crossover digital pre distortion (CO-DPD) technique was proposed. CO-DPD takes into consideration mutual coupling in the DPD formulation. The CO-DPD technique, however, requires substantial computational resources and power, and DPD techniques may be replaced by these new techniques. Replacing DPD techniques with these new techniques requires huge design effort, and may not be cost effective.